Audio power amplifiers of conventional design suffer from low efficiency, and this causes these designs to generate heat that must be removed by large heat sinks, causing the physical amplifier designs to be quite large. In order to make amplifiers smaller, high-efficiency designs have been proposed. Switching amplifiers, also known as class-D amplifiers, are analogous to switching regulators, and so have similar advantage, when compared to class-AB amplifiers, in efficiency and its derivatives, i.e., lower thermal dissipation, longer battery life, smaller power supplies, size, weight, etc. These amplifiers work by converting the analog or digital input signal into a 2-level output signal using a high-frequency modulation process. This 2-level signal is then fed to a power stage to switch the power switches of either a full H-bridge or a half H-bridge. These power switches operate with low switching loss, and thus significantly improve the efficiency of the amplifiers. For switching amplifiers, most prior-art systems employ a pulse width modulation (PWM) scheme. A typical audio PWM amplifier can work at a switching frequency of between 100 KHz ad 500 KHz. Higher switching frequencies will reduce distortion but also result in lower efficiency due to the extra transitions in the output waveform.
FIG. 1 is a circuit diagram of a conventional audio PWM switching amplifier, which includes a full H-bridge 10 and a controller 12. The H-bridge 10 is constructed by four power switches M1-M4 connected between a power supply Vdd and a ground terminal GND. The controller 12 provides PWM signals PWM_P and PWM_N according to an input signal Vin, to switch the power switches M1-M4 so as to generate a differential output voltage OUT=OUTP−OUTN between two output terminals OUTP and OUTN, which is filtered by a low-pass filter (LPF) 14 to filter out audio components contained therein before being applied to a load 16. In further detail, FIG. 2 is a circuit diagram of a conventional PWM generator, in which a comparator CMP compares the input signal Vin with a trianglewave signal Vt to generate the PWM signals PWM_P and PWM_N in opposite phases to each other. Hence, as shown in FIG. 3, the voltage at the positive output terminal OUTP switches between 0 and Vdd, the voltage at the negative output terminal OUTN is in opposite phase to OUTP, and the differential output voltage OUT switches between Vdd and −Vdd. The load current IL decays in the inductors L1 and L2 during the time period of the differential output voltage OUT at −Vdd, and re-establishes in the opposite direction during the period of the differential output voltage OUT at Vdd. Since the output squarewave OUT has the amplitude of 2Vdd and has the duty of 50% when the input signal Vin is zero, the load current IL will have a great ripple and the equivalent series resistance (ESR) of the LPF 14 will cause a great power consumption.
On the other hand, the output filter 14 will reduce the efficiency of the switching amplifier and typically includes external inductors L1, L2 and capacitors C1-C3 which are expensive and consume undesirable amounts of space. Therefore, filterless switching amplifiers have been proposed. However, this will require that the load 16 be inductive. Considering a pure resistive load 16, switching the H-bridge 10 in a binary fashion would place the power supply voltage Vdd across the load 16. Unlike the current waveform IL shown in FIG. 3, the resulting load current IL would be a squarewave with a magnitude equal to the power supply voltage Vdd divided by the resistance of the load 16, and this is with no signal. Although the electrical equivalent of a speaker is somewhere between purely resistive and purely inductive, this would still prevent filterless switching amplifiers in audio applications as the main benefit of efficiency is lost. Even in case of inductive loads, for operation near zero crossing, or no audio signal (Vin=0), the majority of the load current IL is wasted, and is a drop in efficiency, in addition that high electro-magnetic interference (EMI) is produced. Disclosed in U.S. Pat. Nos. 6,262,632 and 6,211,728 is a filterless switching amplifier having a ternary modulation scheme implemented in an H-bridge configuration to eliminate the zero-input load current IL, which operates the H-bridge 10 with a common mode for zero crossing state, by which the two opposite terminals OUTP, OUTN of the load 16 are simultaneously switched between the power supply Vdd and the ground terminal GND. In further detail, at zero input, the H-bridge 10 is switched between two states, one is that the transistors M1, M3 are both turned on to apply the power supply voltage Vdd to the output terminals OUTN, OUTP, and the other is that the transistors M2, M4 are both turned on to ground the output terminals OUTN, OUTP. Consequently, there is no current wasted at zero input. However, the common mode voltage bounces between 0 and Vdd, and this would still cause severe EMI.
Disclosed in U.S. Pat. Nos. 6,847,257 and 6,970,123 are apparatus and methods to reduce the output voltage amplitude and the common mode voltage difference in filterless switching amplifiers. FIG. 4 is a circuit diagram showing the PWM generator used therein, in which, in addition to a comparator CMP to compare the input signal Vin with a trianglewave signal Vt to generate the positive PWM signal PWM_P, another comparator CMP1 is used to compare a signal Vin1 which is an inverse to the input signal Vin, with the same trianglewave signal Vt to generate the negative PWM signal PWM_N. FIG. 5. is waveform diagram showing the differential pair Vin, Vin1 and illustrating how they determine the waveform of the PWM signals OUTP, OUTN. This method reduces the output waveform amplitude OUT of the switching amplifier to Vdd and thus reduces the EMI. However, a minimum pulse width is required for the common mode operation, in order to maintain a non-zero load current at zero input.